1. Field of the Invention
The present invention relates to a wideband low noise amplifier (LNA) having a high dynamic range and used as an RF amplifier for wireless communication devices, and as an input amplifier for A/D converters.
2. Description of the Related Art
An example of conventional low noise amplifiers is a cascode type amplifier disclosed in Unexamined Japanese Patent Application KOKAI Publication No. 2003-289226. The cascode type amplifier has been known as a type of circuit which is not likely to be affected by the parasitic capacitance of a transistor, and which is suitable for an application to a wideband amplifier.
On the other hand, disclosed in “HF Low Noise Amplifiers with Integrated Transformer Feedback”, ISCAS 2002, vol. 2, pp. II-815 to II-818, May, 2002, written by K. van Hartingsveldt, M. H. L. Kouwenhoven, C. J. M. Verhoeven, and A. N. Burghartz is a low noise amplifier circuit having a double-loop negative feedback circuit constituted by a transformer and a resistor. The low noise amplifier circuit having the double-loop negative feedback circuit is a superior circuit which accomplishes a low noise factor, a stable gain, and a good input impedance matching at the same time in a wideband.
It is possible in principle to realize a low noise amplifier having a high dynamic range and operating with low power consumption by causing a Transformer Feedback Cascode LNA (hereinafter, simply called TFC-LNA), constituted by combining the foregoing cascode type low noise amplifier with the foregoing double-loop negative feedback scheme circuit by a transformer and a resistor, to have a high feedback loop gain. However, acquiring a high feedback loop gain and increasing the cut-off frequency of a feedback loop gain transfer function are in a trade-off relationship. Accordingly, if attempting to maintain a high feedback loop gain at a high frequency band, a sufficient phase compensation cannot be achieved in case of applying a conventional compensation method, and oscillation is easily caused, and such an amplifier does not work as an amplifier itself.
Example cases where a conventional phase compensation method is applied to prevent oscillation of the TFC-LNA are a dominant pole compensation method (first conventional example) and a Miller compensation method (second conventional example), both of which are generally used phase compensation methods, and explanations will be given of the respective characteristics thereof.
FIG. 14 is a circuit diagram showing an example of a TFC-LNA 10A where a dominant pole compensation method is applied (first conventional example).
According to the first conventional example, a 10-V direct-current supply voltage Vd1 is applied from a direct-current voltage source DCS, and a transistor having 8 GHz of a transient frequency is used. A signal source 1 having 50Ω of an output impedance is connected to the hot side of a primary winding of a transformer 3 via a direct-current cut-off capacitor 2. A commercially-available transformer whose turn ratio is, for example, 1:2 is used as the transformer 3.
The cold side of the primary winding of the transformer 3 is connected to the base of an NPN transistor 4. The base of the transistor 4 is connected to the positive electrode of a biasing voltage source 5 via a choke coil 6.
The collector of the transistor 4 is connected to the emitter of an NPN transistor 7. The base of the transistor 7 is connected to the positive electrode of a biasing direct-current voltage source 8, and is grounded from the standpoint of an alternate current. The transistor 4 and the transistor 7 are connected together in a cascode manner, thereby constituting a cascode amplifier having a resistor 9 as a load. The collector of the transistor 7 is connected to one end of the resistor 9 which functions as a load for the cascode amplifier. The direct-current supply voltage Vd1 is applied to the other end of the resistor 9.
The node between the resistor 9 and the collector of the transistor 7 functions as an output node of the cascode amplifier where an amplified output voltage signal is output, and is connected to the base of an NPN transistor 10, i.e., the input terminal of an emitter follower. The transistor 10 and a constant current source 18 constitute the emitter follower, and work as an output buffer of the TFC-LNA 10A. The direct-current supply voltage Vd1 is applied to the collector of the transistor 10 from a direct-current voltage source DCS. A phase compensation capacitor 11 is connected between the base of the transistor 10, i.e., the output node of the cascode amplifier and the positive electrode of the direct-current voltage source DCS, i.e., an alternating current reference potential ground, and the load resistor 9 and the capacitor 11 give a dominant pole of a feedback loop gain, and function as a low-pass filter for the output of the cascode amplifier. The emitter of the transistor 10, i.e., the output terminal of the TFC-LNA 10A, is connected to a load 13 of that amplifier via a direct-current cut-off capacitor 12. In the example of FIG. 14, the load 13 comprises a resistor of 5 kΩ.
The cold side of the secondary winding of the transformer 3 is connected to the emitter of the transistor 10, i.e., the output terminal of the TFC-LNA 10A. The hot side of the secondary winding is connected to the ground. An output voltage signal applied to the secondary winding of the transformer 3 is transmitted to the primary side of the transformer 3 by electromagnetic coupling, and is series-mixed with an input signal. This constitutes the first negative feedback path of the TFC-LNA 10A. A resistor 16 and a direct-current cut-off capacitor 17 are connected in series between the emitter of the transistor 10, i.e., the output terminal of the TFC-LNA 10A, and the hot-side terminal of the primary winding of the transformer 3, i.e., the signal input terminal of the TFC-LNA 10A, and function in such a way that an output signal is shunt-mixed with an input signal. This constitutes a second negative feedback path of the TFC-LNA 10A. The emitter of the transistor 10 is connected to, for example, a constant current source 18 to supply an operating current of the emitter follower.
According to the first conventional example, the operating current of the emitter follower is set to approximately 12 mA. Because the cascode amplifier with a voltage gain of 200 (46 dB) is used in the TFC-LNA 10A of the first example, the maximum magnitude of its feedback loop gain becomes over 40 dB.
The voltage gain of the TFC-LNA 10A is theoretically given by the turn ratio N of the transformer 3, and because the turn ratio of the commercially-available transformer used in the first conventional example is 1:2, the voltage gain of the TFC-LNA 10A of the first conventional example is approximately 6 dB. The commercially-available transformer 3 used in the first conventional example is a transformer having a loss of 1.0 dB or so and a pass band between 3 MHz to 200 MHz. The most appropriate resistance of the resistor 16 which serves as a feedback resistor is theoretically given by an equation (N+1)R, where R is an input impedance set as a specification of the TFC-LNA 10A and N is a turn ratio of the transformer 3. According to the first conventional example, the input impedance R is set to 50Ω which is a general value, and the resistance of the resistor 16 is 150Ω.
As explained above, the load resistor 9 and the capacitor 11 function as to cause a dominant pole in the transfer function of a feedback loop gain, and the effect thereof results in a phase compensation of the TFC-LNA 10A of the first conventional example. Regarding the TFC-LNA 10A of the first conventional example, in a case where a phase compensation is carried out in such a way that the feedback loop gain measured at the base of the transistor 4 has a phase margin of about 45°, it is necessary for the capacitor 11 to have a capacitance of greater than or equal to 140 pF. It is difficult to form such a large-capacitance capacitor on an integrated circuit in view of a cost limitation, and such capacitor must be an external part. This causes demerits like increments of a part number and a substrate area, and it is a disadvantage of a dominant pole compensation method.
FIG. 15 is a diagram showing the feedback loop gain of the TFC-LNA 10A of the first conventional example in Bode plotting, where a result of measuring the feedback loop gain measured at the base of the transistor 4 in simulation is plotted.
The feedback loop gain of the TFC-LNA 10A of the first conventional example gives the maximum value of about 44 dB around a frequency of 360 kHz, and decreases to 0 dB around a frequency of about 190 MHz. The phase margin is 45°, and the gain margin is about 5 dB. The −3 dB cut-off frequency which indicates a frequency where the feedback loop gain starts decreasing is about 1.1 MHz, and thus it becomes apparent that the band where the TFC-LNA 10A maintains a high dynamic range is merely several MHz or so.
According to the dominant pole compensation method, a compensation is carried out in such a way that the feedback loop gain decreases at −20 dB/dec as the frequency of an input signal increases. The maximum frequency of the pass band where the commercially-available transformer used in the first conventional example works substantially ideally is about 200 MHz, and at a frequency band higher than this, the deterioration of the phase margin due to the parasitic capacitance or the like of the transistor becomes notable, in addition to the attenuation of the phase margin due to dominant pole.
Accordingly, if attempting to have a sufficient phase margin through the foregoing method, it is necessary to set the cut-off frequency low to 200 MHz/2 dec (=100), i.e., less than or equal to 2 MHz in a case where the maximum feedback loop gain is set to a high value greater than or equal to 40 dB like the first conventional example. In this manner, the band where the high feedback loop gain is maintained is limited to a frequency remarkably lower than the transformer-pass-band maximum limit, and the TFC-LNA 10A does not bring out a sufficient performance when used as a high-frequency low noise amplifier. This is another disadvantage when the dominant pole compensation method is used.
FIG. 16 is a diagram showing a simulation result of measuring the third order input intercept point characteristic for the TFC-LNA 10A of the first conventional example (FIG. 14).
In the simulation of the third order input intercept point (hereinafter, IIP3) of the TFC-LNA 10A of the first conventional example (FIG. 14), two tone signals each having power of −50 dBm are used as inputs at a frequency shifted by ±10 kHz around the measured frequency. The axis of abscissas in FIG. 16 represents a frequency (MHz), and the axis of ordinates thereof represents IIP3 (dBm).
As is apparent from FIG. 16, IIP3 deteriorates greater than or equal to 20 dB from the maximum value of 42 dB at 10 MHz. The deterioration of IIP3 is caused in response to the attenuation of the feedback loop gain of the TFC-LNA 10A shown in FIG. 15. In general, according to a negative feedback amplifier, as the feedback loop gain thereof decreases, the value of IIP3 decreases together. As explained above, according to the TFC-LNA 10A where the dominant pole compensation method is applied, it is difficult to maintain a high feedback loop gain at a high frequency band, so that the circuit of the first conventional example is not suitable for a high-frequency low noise amplifier that a good distortion characteristic, i.e., a high dynamic range is required.
Next, an explanation will be given of an example case where a phase compensation using a Miller compensation method is applied to a TFC-LNA as a second conventional example.
FIG. 17 is a circuit diagram showing an example where a phase compensation using a Miller compensation method is applied to a TFC-LNA (second conventional example).
A TFC-LNA 10B of the second conventional example is a circuit where a Miller compensation method is applied to a TFC-LNA, having the same specifications (direct-current supply voltage Vd1=10 V, voltage gain=6 dB, and input impedance=50Ω) as those of the TFC-LNA 10A of the first conventional example, in lieu of a dominant pole compensation method. That is, in the TFC-LNA 10B of the second conventional example, all elements other than a phase compensation capacitor 19, and voltage sources and a current source have common circuit constants to those of the TFC-LNA 10A of the first conventional example, and the same transistor having a transient frequency of 8 GHz is used.
While according to the first conventional example, the phase compensation capacitor 11 is connected between the output node of the cascode amplifier, i.e., the collector of the transistor 7, and the alternating current ground, according to the second conventional example, however, the phase compensation capacitor 19 is connected between the output node of the cascode amplifier and the input node of the cascode amplifier, i.e., the base of the transistor 4. Let us suppose that the capacitance of the phase compensation capacitor 19 is C and the voltage amplification degree of the cascode amplifier is β, then, the phase compensation capacitor 19 connected in a Miller compensation manner functions as to have an approximately same effectiveness as that of a case where a shunt capacitor having a capacitance of (β−1)C is connected to the input node of the cascode amplifier. Accordingly, in general, in the Miller compensation method, it is possible to perform a phase compensation using a capacitor of a smaller capacitance in comparison with the dominant pole compensation method.
FIG. 18 is a diagram where the feedback loop gain of the TFC-LNA 10B of the second conventional example is indicated in Bode plotting, and where a result of measuring the feedback loop gain measured at the base of the transistor 4 in simulation is plotted.
As shown in FIG. 18, the decrement of the loop gain becomes suddenly gradual from around 550 MHz. This indicates that the TFC-LNA 10B starts operating in such a way that the capacitor 19 feeds forward a signal around that frequency, i.e., a signal applied to the input node of the cascode amplifier is transmitted to the base of the emitter follower transistor 10 passing through the capacitor 19. This corresponds to the fact that a zero is formed in the input-output transfer function of the TFC-LNA 10B, due to the presence of an additional signal path from the input to the output through the capacitor 19. Accordingly, the cross over frequency of the feedback loop gain becomes higher, making it difficult to maintain the phase margin greater than 0°. Furthermore, because the feedback loop gain attenuates at a sharp inclination of about −60 dB/dec between 200 to 500 MHz, the phase attenuation inclination becomes precipitous around this band.
In a case where the capacitance of the capacitor 19 is increased to a further larger value, and the cut-off frequency of the feedback loop gain is lowered to do phase compensation, the capacitor 19 starts operating in the feed forward direction at a frequency lower than 550 MHz, and the attenuation inclination of the feedback loop gain becomes gradual, so that the cross point frequency does not sufficiently decreases, resulting in destabilization of the TFC-LNA. In other words, it is difficult to obtain sufficient phase margin by increasing the capacitance of the capacitor 19 so that the cut-off frequency of the feedback loop gain is lowered enough, because the frequency of the zero in the input-output transfer function caused by the capacitor 19 is also lowered simultaneously, offsetting the decrease of the cross over frequency of the feedback loop gain. It is possible to prevent the cross over frequency from becoming high by decreasing the feedback loop gain, but the IIP3 characteristic of the amplifier also deteriorates simultaneously due to the reduction of the feedback amount of the amplifier. Because of the foregoing reason, it is difficult to realize a phase compensation for a TFC-LNA having a high dynamic range by the Miller compensation method.